Introduction of Power Switching Devices-Characteristics
Introduction of Power Switching Devices-Characteristics
In Power Electronic Systems (PES), the most important feature is the efficiency.
Therefore as a rule PES do not use resistance as power circuit elements.
The function of dropping voltages and passing currents is therefore, achieved by means of switches.
The ideal switch drops no voltage (zero resistance) while ON and passes no current (zero conductance) while OFF.
When a switch is operated alternately between the two (ON and OFF) states, it may be considered to oer an eective resistance depending on the switching duty ratio.
Eectively the switch functions as a loss-less resistance.
However, the load voltage and current are not smooth on account of the switching process in the control.
In general PES will consist of switches for the control.
Power flow and reactive elements (filters) to divert the eects of switching from reaching the load.
The power circuit elements in PES are therefore
✓Switches - (to control transfer of energy)
✓Reactors - (Inductors and Capacitors) (to smoothen the transfer of energy)
Ideal Switches
There are several electronic devices, which serve as switches.
We may at first list out the desired features of ideal switches.
The practical devices may then be studied with reference to these ideal characteristics.
The features of ideal switches (with reference to the schematic shown )
Diagram
✓ In the OFF state, the current passing through the switch is zero and the switch is capable of supporting any voltage across it.
✓In the ON state, the voltage across the switch is zero and the switch is capable of passing any current through it.The power dissipated in the switch in the ON and OFF states is zero.
✓The switch can be turned ON and OFF instantaneously.
✓The switch does not need energy to switch ON/OFF or OFF/ON or to be maintained in the ON/OFF states.
✓The switch characteristics are stable under all ambient conditions.
Features 1 and 2 lead to zero conduction and blocking losses.
Feature 3 leads to zero switching losses. Feature 4 leads to zero control eort.
Feature 5 makes the ideal switch indestructible.
The operating points of the ideal switch on the VI plane lie along the axis as shown in Fig. 4.
Practical devices, though not ideal, reach quite close to the characteristics of ideal switches.
Real Switches
Real switches suer from limitations on almost all the features of the ideal switches.
✓The OFF state current is nonzero. This current is referred to as the leakage current. The OFF state voltage blocking capacity is limited.
✓The ON state voltage is nonzero. This voltage is called the conduction drop. The ON state current carrying capacity is limited. There is nite power dissipation in the OFF state (blocking loss) and ON state (conduction loss).
✓Switching from one state to the other takes a nite time. Consequently the maximum operating frequency of the switch is limited. The consequence of nite switching time is the associated switching losses.
✓The switch transitions require external energy and so also the switch states. Real switches need supporting circuits (drive circuits) to provide this energy.
✓The switch characteristics are thermally limited. The power dissipation in the device is nonzero. It appears as heat and raises the temperature of the device.
✓To prevent unlimited rise in temperature of the device external aids are needed to carry away the generated heat from the device.
✓Real switches suer from a number of failure modes associated with the OFF state voltage and ON state current limits.
✓The operating points of real switches on the VI plane are shown in Fig.1.5.
✓The steady state operating points lie close to the axis within certain limits.
✓Further there is a safe operating area (SOA) on the VI plane for transient operation.
Practical Power Switching Devices
There are several power switching devices available for use in PES. They may be classied as,
A Uncontrolled switches
The state (ON/OFF) of the switch is determined by the state of the power circuit in which the device is connected. There is no control input to the device. Diodes are uncontrolled switches.
B Semi-controlled switches
The switch may be turned to one of its states (OFF/ON) by suitable control input to its control terminal. The other state of the switch is reachable only through intervention from the power circuit.
A thyristor is an example of this type of switch. It may be turned ON by a current injected into its gate terminal; but turning OFF a conducting thyristor is possible only by reducing the main current through the device to zero.
C Controlled switches
Both the states of the switch (ON/OFF) are reachable through appropriate control signals applied to the control terminal of the device. Bipolar junction transistor (BJT), eld eect transistor (FET), gate turn-off thyristor (GTO), insulated gate bipolar transistor (IGBT) fall under this group of switches.
The switches desired in PES are realized through a combination of the above devices.
Diodes
The diode is a two terminal device - with anode (A) and cathode (K). The v-i characteristic of the diode is shown in Fig. 1.6
✓When the diode is forward biased (VAK > 0), the diode approximates to an ON switch.
✓When the diode is reverse biased (VAK < 0), the diode approximates to an OFF switch.
For a typical application, the forward and reverse biased operating points are shown in Fig. 1.7.
✓In the ON and OFF condition, the diode dissipates certain nite power.
✓The diode does not have explicit control inputs. It reaches the ON state with a small delay (tr) when the device is forward biased. It blocks to the OFF state after a small delay (trr) when the forward current goes to zero
✓The forward recovery time is much less than the reverse recovery time.
During the reverse recovery time a negative current flows through the device to supply the reverse charge required to block reverse voltage across the junction.
The process is shown in Fig.1.8. The reverse recovery time decides the maximum frequency at which the diode may be switched.
Thyristor or Silicon Controlled Rectier (SCR)
The Thyristor is a four-layer device. It has three terminals - anode (A), cathode (K), and gate (G). The anode and the cathode form the power terminal pair. The gate and cathode form the control terminal pair. The characteristic of the SCR without any control input is shown in Fig. 9.
Under forward biased condition, junction 2 (J2) supports the entire voltage. Under reverse biased condition the junction 1 (J1) supports the entire voltage. Junction 3 (J3) is the control junction and cannot support appreciable reverse voltage.
The reverse and forward blocking currents for the SCR are of the same order (a few mA). The control action of the SCR is best understood from the classical two-transistor model shown in Fig. 10. The anode current in the model may be written as
where α1 and α2 are the common base current gains of the transistors and Icbo1 and Icbo2 are their leakage currents; α1 and α2 are low at low anode currents.
In the absence of control (IG = 0), the anode current will be a small leakage current. There are several mechanisms by which the SCR may be triggered into conduction.
Gate turn-on
If gate current IG is injected, the emitter currents of the component transistors increase by normal transistor action.
The device switches regeneratively into conduction when IG is suffciently high. Once the device turns ON, the gate circuit has no further influence on the state of the SCR.
Voltage turn-on
If the forward anode blocking voltage is slowly increased to a high value, the minority carrier leakage current across the middle junction increases due to avalanche eect.
This current is applied by the transistor action leading to eventual turn-on of the SCR.
dV/dt turn-on
When the anode voltage rises at a certain rate, the depletion layer capacitance of the middle junction will pass a displacement current (i = CdV=dt).
This current in turn will be amplified by transistor action leading to the turn-on of the device.
Temperature effect
At high junction temperature, the leakage currents of the component transistors increase leading to eventual turn-on of the device.
Light ring
Direct light radiation into the gate emitter junction will release electron-hole pairs in the semiconductor.
These charge carriers under the influence of the across the junction will flow across the junction leading to the turn-on of the device.
Switching Characteristics of the SCR
The switching operation of an SCR is shown in Fig. 1.12. The important features are
Initially when forward voltage is applied across the device, the off state or static dV/dt has to be limited so that the device does not turn ON.
When gate current is applied (with anode in forward blocking state), there is a nite delay time before the anode current starts building up. This delay time 'td", is usually a fraction of a microsecond.
After the delay time, the device conducts and the anode current builds up to the full value IT . The rate of rise of anode current during this time depends upon the external load circuit.
If during turn-on, the anode current builds up too fast, the device may get damaged. The initial turn-on of the device occurs near the gate cathode periphery and then the turn-on area of the device spreads across the entire junction with a nite velocity.
If IT rises at a rate faster than the spreading velocity, then the entire current IT is conned to a small area of the device eventually causing overheating of the junction and destruction of the device.
Therefore it is necessary to limit the turn-on di/dt of the circuit to less than the safe di/dt that can be tolerated by the device.
During conduction, the middle junction is heavily saturated with minority carriers and the gate has no further control on the device. The device drop under this condition is typically about 1V.
From the conducting state, the SCR can be turned OFF by temporarily applying a negative voltage across the device from the external circuit. When reverse voltage is applied, the forward current rst goes to zero and then the current builds up in the reverse direction with the commutation di/dt. The commutation di/dt depends on the external commutating circuit.
The reverse current flows across the device to sweep the minority carriers across the junction. At maximum reverse recovery current IRM, the junction begins to block causing decay of reverse current. The fast decay of the recovery current causes a voltage overshoot VRRM across the device on account of the parasitic inductance in the circuit.
At zero current, the middle junction is still forward biased and the minority carriers in the vicinity must be given time for recombination. The device requires a minimum turn-on time tq before forward blocking voltage may be applied to the device. The reapplied dV/dt has to be limited so that no spurious turn-on occurs. The device turn-off time 'tq" is a function of
Thyristors are available for PES applications with voltage ratings upto about 3000V and current ratings upto about 2000A.
✓When blocking forward or reverse voltages a small leakage current flows.
✓When conducting forward current a low voltage is dropped.
✓There are nite power losses in conduction and blocking.
✓The turn ON and turn OFF processes are not instantaneous.
✓The device losses warrant proper thermal design.
✓Turn-ON requires energy through gate circuit. Usually this is quite small.
✓Turn-OFF requires energy supplied through an external commutation circuit. This energy usually is much larger than the turn-on energy supplied through the gate.
✓SCR passes unidirectional current and blocks bi-directional voltage.
Bipolar Junction Transistor (BJT)
The transistor is a three terminal device - emitter (E), collector (C), and base (B). The collector-emitter forms the power terminal pair. The base-emitter form the control terminal pair.
The vi characteristics of the transistor is shown in Fig. 1.13. There are three distinct regions of operation. In the cut-off region, the base current is zero and the device is capable of blocking forward voltage. In the active region, the collector current is determined by the base current (ic = βib). In the active region of operation the device dissipation is high.
In the saturated region, the base is overdriven (ib ≥ ic/β). The device drops a small forward voltage and the current is determined by the external circuit. When used as a switch, the transistor is operated in the cut-off and saturated regions to achieve OFF and ON states respectively.
In the cut-off region (OFF state), both base-emitter and base-collector junctions are reverse biased. In the saturated region of operation (ON state), both base-emitter and base-collector junctions are forward biased (ib≥ic/β). The features of the transistor in switching applications are:
✓The conduction, blocking and switching losses raise the junction temperature of the device. To limit the operating junction temperature of the device, proper thermal design has to be made.
✓The device requires drive circuits.
✓The transistor blocks positive voltage and passes positive current.
✓During transients (OFF/ON and ON/OFF), the operating point of the switch requires to be limited to stay within the safe operating area (SOA) of the v-i plane.
Switching Characteristics of the Transistor
The switching performance of the transistor is shown in Fig. 14. The important features are
Turn On
The base gets charged.
After a delay of “td", the collector junction starts conduction.
In a time “tr", the collector-emitter voltage drops (almost) linearly to Vce(sat).
The collector current starts from the moment the collector-emitter voltage starts falling. The rise of collector current with time during (hatched region 1) this transient is decided by the external circuit.
Turn off
To turn-off the transistor, the forward base drive is removed and a negative base drive is set up. The junctions (base-emitter and base-collector) remain forward biased for a duration "ts".
During this storage time, ic continues to flow and the device voltage vce drop remains low. This is the time taken to remove the accumulated charge in the junction, so that the junction may start blocking.
The storage time increases with ib1 and decreases with ib2. After the storage time, in a time "tf ", the collector current falls (almost) linearly to zero. During the fall time (hatched region 2), the collector emitter voltage vce(t) is decided by the external circuit.
It may be seen from the switching process that the device losses are low during the transient intervals td and ts. The switching losses occur during tr and tf .
The collector current during tr and the device voltage during tf are dictated by the external circuit. This feature is used to reduce the switching losses in any application.
The important specification of the transistor are
Peak and average current (to assess suitability with a power circuit)
Peak blocking voltage Vceo (to assess suitability with a power circuit)
ON state voltage Vce(sat) (to assess conduction loss)
OFF state current Iceo (to assess blocking loss)
Thermal impedance (to help thermal design)
Switching times td, ts, tr and tf (to design drive circuits Ib1, Ib2 and to select the switching frequency)
Forced beta (to design drive circuit)
Safe operating area SOA (to design switching protection)
MOS Field Effect Transistor(MOSFET)
MOSFET is becoming popular for PES applications at low power and high frequency switching applications (a few kW and a few 100s of kHz). It is a three terminal device - drain (D), source (S) and gate (G).
Drain and source form the power terminal pair. Source and gate form the control terminal pair. The gate is insulated from the rest of the device and therefore draws no steady state current.
When the gate is charged to a suitable potential with respect to the source, a conducting path known as the channel is established between the drain and the source.
Current ow then becomes possible across the drain and the source. MOSFETs used in PES are of enhancement type, i.e. the device conducts when a suitable gate to source voltage is applied. Whenever the gate to source voltage is zero, the device blocks. Both N and P channel MOSFETs are available.
The N channel versions are more common. The v-i characteristic of a MOSFET is shown in Fig. 15. The two regions of operation are the cut-o region (OFF state) when Vgs is 0 and the resistance region (ON state) when Vgs is greater than Vgs(th).
The device has no reverse blocking capability on account of the body diode, which conducts the reverse current.
The features of the MOSFET in switching applications are,
Turn-off
The turn-off process in the GTO is initiated by passing a negative current through the gate cathode circuit. The cathode current is then constricted towards the centre of each cathode segment, thus pinching off the cathode current. As the cathode current is pinched, the anode current falls rapidly. During the pinch- off process the active silicon area reduces. Further, the cathode current tends to get redistributed away from the extinguishing gate current. This process takes place during the storage time. This process culminates with a rising anode voltage and a falling anode current. This phase is the most critical in the turn-off process and requires the presence of a snubber across the device to limit the reapplied rate-of-rise-of-anode-voltage to about 500 to 1000 v/s. This process is shown in Fig. 1.18. The GTO zone in Fig. 1.18 is a vulnerable zone when both anode voltage and cathode current co-exist. In order to prevent the device from turning on again in this region, it is necessary, to use an appropriate snubber across the device
Blocking
In the blocking state, the GTO behaves just like a PNP transistor. When the bias supply has negligible impedance, the GTO has practically unlimited dv/dt capability
Gate Drive
The gate drive for the GTO has the following requirements.
Turn the GTO on by means of a high current pulse (IGM).
Maintain conduction through provision of a continuous gate current during on state.
Turn-off the GTO with a high negative current pulse (IGQ).
Maintain negative gate voltage during o state with suciently low impedance.
Typical gate drive waveforms are shown in Fig. 19. GTOs require much more gate current than a similar rated SCR. The GTO structure is well suited for high-current pulsed applications on account of their large turn-on rate-ofrise- of-anode-current capability. The initial gate current IGM and the recommended value of dIG=dt can be taken from the data sheet. A rough guide to the required value of IGM is that it is about 6 times IGT . For a GTO with 3A IGT at 25oC, IGM is 20A at 25oC, or 60A at -40oC, for the values of anode voltage and di/dt cited on the data sheet (50% of Vdrm and 300 to 500/μS).The rate of rise of this current is important. It should be atleast 5% of the anode di/dt, with a minimum duration equal to the sum of the delay time and rise time (tgt = td tr).
In the conduction state, the GTO is like a Thyristor. Extra care must be taken such that the GTO does not partially unlatch following turn-on. This may happen in motor drive applications, where the load current may fall momentarily to a low value following turn-on. The continuous drive current is to overcome such eventualities. This current must be atleast 20% more than IGT . This is all the more important when the load current becomes negative, when the load current
ows through the freewheeling diode. In such a case the GTO returns to the off state. Then when the load current becomes positive, the GTO will not turn on. A typical 3000A GTO requires a continuous drive current of the order of 10A at -40C. This need is highly temperature dependent. The turn-off of the GTO requires atleast 20% to 30% of anode current. Fig. 1.18 shows a critical period during which the anode voltage is positive and the cathode current is non-zero. The drive design must provide a large dIGN=dt, to minimize this critical duration. In the blocking state, the preferred gate bias is about -5V or lower. This may be as high as the rated gate-cathode voltage. Under this situation, the device has practically no dv/dt limitation. The device behaves as a low gain BJT with open base.
Switching Characteristics of the IGBT
The switching performance of the IGBT is shown in Fig. 1.21. The switching process may be seen to be a combination of the switching performance of a
MOSFET and a BJT.
Just as in a transistor, the current rise in region 1 and the voltage build up in the region 2 are determined by the external circuit.
The important specifications of the IGBT are
Peak and average current (to assess suitability with a power circuit)
Peak blocking voltage Vces (to assess suitability with a power circuit
ON state voltage Vce(sat) (to assess conduction loss)
OFF stage current Ices (to assess blocking loss)
Thermal impedance (to help thermal design)
Switching times td; tr; ts; tf
(to design drive circuit and to select switching frequency)
Threshold voltage Vge(th) (to design drive circuit)
Safe operating areas SOA (to design switching protection
Integrated Gate Commutated Thyristor (IGCT)
Developed in 1994 and announced in 1997, the Integrated Gate-Commutated Thyristor is the latest addition to the thyristor family. It combines the rugged on state performance of the thyristors and the positive features of the turn-off behaviour of the transistor. The Gate-Commutated Thyristor is a semiconductor based on the GTO structure, whose gate circuit is of such low inductance that the cathode emitter can be shut off instantaneously, thereby converting the device during turn-off to effectively a bipolar transistor. The basic principle of operation of the IGCT is illustrated in Fig. 1.22. In the conducting state the IGCT is a regenerative thyristor switch. It is characterized by high current capability and low on-state voltage. In the blocking state, the gate-cathode junction is reverse-biased and is selectively out of operation. The equivalent,
circuit of the blocking state is as shown in Fig. 1.23. Fig. 1.22 is identical to the conducting and blocking states of GTOs. The major dierence with IGCT is that the device can transit from conducting state to blocking state instantaneously. The GTO does so via an intermediate state as illustrated in Fig. 1.18. In IGCT technology, elimination of the GTO zone is achieved by quickly diverting the entire anode current away from the cathode and out of the gate. The device becomes a transistor prior to it having to withstand any blocking voltage at all. Turn-o occurs after the device has become a transistor, no external dv/dt protection is required. IGCT may be operated without snubber like IGBT or MOSFET. Fig. 1.24 shows the turn-o process of an IGCT. Notice,
Thermal Design of Power Switching Devices
The cross section of a power switching device (a diode) and its thermal model are shown in Fig. 1.26. We have seen that the real power switching devices
dissipate energy unlike the ideal switching devices. These losses that take place in the device are
Conduction loss
Blocking loss
Turn-on loss
Turn-off loss
All these energy losses in the device originate at the junction. Unless these losses are carried away from the junction, the temperature of the device junction will rise without limit and eventually destroy the device. In this section we see the basics of the thermal process in the device. The thermal model thus established may be used to design heat sinks for the device to limit the temperature rise of the device junction. Part of this heat generated at the junction increases the temperature of the junction and the rest flows out of the junction onto the case of the device and therefrom to the environment of the device.
Thermal model of the device
Let P(t) be the power dissipated in the junction. Let the initial temperature of the junction and the case be θj(0) and θc(0) respectively in oC. In time "dt" let the increase in temperature of the junction and the case be dθj(t) and dθc(t) respectively in C. J is the caloric equivalent of joule. For heat balance in time dt,
PJdt = Heat generated in the junction in time dt
msdθj = Heat retained in the junction in time dt
m = mass of the semiconductor material in Kg
s = specic heat of the junction material Cal/Kg/oC.
Kθ[θj(t) - θc(t)] = Heat taken away from the junction to case in time dt
K = thermal conductivity from junction to case in Cal/oC/sec
The above heat balance equation may be rearranged as follows,
For the purpose of simple analysis we may assume that the case temperature θc(t) to be constant at c . We may further dene a new variable as the temperature dierence between the junction and the case θjr(t).In the new variable ,
Cth = Thermal Capacity of the Junction in J=oC or J=oK:
Rth = Thermal Resistance of the Junction in oC=W or oK=W:
The above dierential equation relates the thermal behaviour of the junction, and when solved will give the junction temperature rise as a function of time.
Transient temperature rise
Under transient conditions the dierential equation of the thermal model has to be solved to obtain the temperature rise as a function of time. However, if
we consider transients of very small duration (as happens in power switches during switching), P(t) may be considered constant. Then θjr, may be solved
Intelligent Power Modules (IPM)
The introduction of MOS technology in the fabrication of power semiconductors has created great device and application advantages. Of particular interest is the current modern power device namely the insulated gate bipolar transistor (IGBT). Currently IGBTs are taking several applications away from MOSFET modules at the low power high frequency end and from bipolar, Darlington modules at the high power medium frequency end of the application spectrum. With such technology, it has become possible to integrate the peripheral devices to be built into the power modules. Such devices are classified as Intelligent Power Modules (IPM). The dierent levels of integration achieved and achievable are shown in Fig. 1.29. The IPM family provides the,user with the additional benets of equipment miniaturization and reduced design cycle time. They include gate drive circuit and protection circuits for short circuit protection, over-current protection, over-temperature protection, and gate drive under-voltage lockout. IPM has sophisticated built-in protection circuits that prevent the power devices from being damaged in case of system malfunction or overload. Control supply under-voltage, over-temperature, over-current, and short-circuit protection are all provided by the IPMs internal gate control circuits. A fault output signal is provided to alert the system controller if any of the protection circuits are activated. Fig. 1.30 is a block diagram of the IPM's internally integrated functions. This diagram also shows the isolated interface circuits and isolated control power supply that must be provided to the IPM. The IPM's internal control circuit operates from an iso-lated 15V DC supply. If for any reason, this voltage falls below the species under-voltage trip level, the power devices will be turned o and a fault signal generated. Small glitches less than the specified tdUV in length will not affect the operation of the control circuit and will be ignored by the under voltage protection circuit. In order for normal operation to resume, the control supply voltage must exceed the under-voltage reset level (UVr). Operation of the under-voltage protection circuit will also occur during power up and power down situation. The system controller must take into account the fault output delay (tfo).
Introduction to Reactive Elements in Power Electronic Systems
The conditioning of power flow in PES is done through the use of electromagnetic and reactive elements (inductors, capacitors and transformers).
In this section the basics of electromagnetics is reviewed.
The type of capacitors popular in power electronic applications are also given.
They are formulated in such a way as to be useful for the design of inductors and transformers.
Electromagnetics
The voltage across and current through a conducting element is related through Ohm's law. This law may be stated as follows. When an electric eld (of intensity V/m) is set up across a conducting material (of conductivity σ1/Ω m), there is an average flow of electrical charge across the conducting material.
(of current density J A/m2). This is shown in Fig. 1.
When expressed in terms of element voltage and current, this reduces to the familiar statement of Ohm's law.
In comparison with conducting materials, the property of magnetic materials may be stated as follows. When a magnetic eld (of intensity H A/m) is set
up across a magnetic material, of magnetic permeability (μ H/m), a magnetic flux of density (B Tesla) is set up in the magnetic material as shown in Fig.2.2.
The above equation, in terms of the magnetomotive force (mmf) F and the,
flux Φ in the magnetic circuit, reduces to,
where, R = reluctance of the magnetic circuit = l=Aμ
The above relationship is analogous to Ohm's law for magnetic circuits. The magnetic permeability of any magnetic material is usually expressed relative
to the permeability of free space . The reluctance of the magnetic circuit is given by,
Electromagnetic circuit elements consist of an electric circuit and a magnetic circuit coupled to each other. The electric current in the electric circuit sets up the magnetic eld in the magnetic circuit with resultant magnetic flux. Seen as an electrical circuit element, the electromagnetic element possesses the property of energy storage without dissipation. Ampere's law and Faradays law,
Design of Inductor
The inductor consists of a magnetic circuit and an electrical circuit. The design requires,
The size of wire to be used for the electric circuit, to carry the rated current safely.
The size and shape of magnetic core to be used such that,
The peak ux is carried safely by the core without saturation.
The required size of the conductors are safely accommodated in the core.
The number of turns of the electric circuit to obtain the desired inductance.
Material constraints
Any given wire (conducting material) can only carry a certain maximum current per unit cross section of the wire size. When this limit is exceeded, the
wire will overheat from the heat generated (I2R) and melt or deteriorate. The safe current density for the conducting material is denoted by J A/m2. Any
magnetic material can only carry a certain maximum flux density. When this limit is exceeded, the material saturates and the relative permeability drops
substantially. This maximum allowable flux density for the magnetic material is denoted by BmT.
Design Relationships
In order to design an inductor of L Henry, capable of carrying an rms current, of Irms and peak current of Ip
The above equation may be interpreted as a relationship between the energy handling capacity (0:5LI2) of the inductor to the size of the core (ACAW), the
material properties (Bm, J), and our manufacturing skill (kw).
kw depends on how well the winding can be accommodated in the window of the core. kw is usually 0.3 to 0.5.
Bm is the maximum unsaturated flux is about 1 T for iron and 0.2 T for ferrites.
J is the maximum allowable current density for the conductor. For copper conductors J is between 2.0x106 A=m2 to 5.0x106 A=m2.
Design steps
Select nearest whole number of N
Design of Transformer
Unlike the inductor, the transformer does not store energy. The transformer consists of more than one winding. Also, in order to keep the magnetization current low, the transformer does not have air gap in its magnetizing circuit. Consider a transformer with a single primary and single secondary as shown in Fig.2.7. Let the specifications be
Primary: V1 volt; I1 ampere;
Secondary: V2 volt; I2 ampere;
VA Rating: V1 I1 = V2 I2;
Frequency: f Hz
For square wave of operation, the voltage of the transformer is,
The window for the transformer accommodates both the primary and the secondary. With the same notation as for inductors,
The above equation relates the area product (ACAW) required for a transformer to handle a given VA rating.
Design Steps
For a given specification of VA, V1, V2, J, Bm, kw, and f, it is desired to design a suitable transformer. The design requires, Size of wire and number of turns to be used for primary and secondary windings. Core to be used.
Resistance of the winding.
Magnetizing inductance of the transformer.
✓Compute the Area product (ACAW) of the desired core.
✓Select the smallest core from the core tables having an area product higher than obtained in step (1).
✓Find the core area (AC) and window area (AW) of the selected core.
✓Compute the number of turns
✓Select the nearest higher whole number to that obtained in step (4), for the primary and secondary turns.
✓Compute the wire size for secondary and primary.
✓Select from the wire tables the desired wire size.
✓Compute the length of secondary and primary turns, from the mean length per turn of the core tables.
✓Find from the wire tables, the primary and secondary resistance.
✓Compute from the core details, the reluctance of the core.
Capacitors for Power Electronic Application
Power electronic systems employ capacitors as power conditioning elements. Unlike in signal conditioning applications, the capacitors in PES are required to handle large power.
As a result they must be capable of carrying large current without overheating. To satisfy the demands in PES, the capacitors must be very close to their ideal characteristics namely low equivalent series resistance (ESR) and low equivalent series inductance (ESL).
Low ESR will ensure low losses in the capacitor. Low ESL will ensure that the capacitor can be used in a large range of operating frequency. Figure 8 shows the impedance of a capacitor as a function of frequency.
It is seen that a real capacitor is
Close to the ideal at lower frequencies. At higher frequencies, the ESR and the ESL of the real capacitor make it deviate from the ideal characteristics.
For PES applications, it is necessary that the ESR and ESL of the capacitor are low.
Base Drive Circuits for BJT
In power electronic applications, the BJTs used will be capable of blocking high voltages (up to about 600V) when OFF, and passing high currents (up
to 50 to 100A) when ON. Such high power transistors are quire sensitive to voltage and current stresses. The high voltage devices are very sensitive to
reverse biased second breakdown. The high current devices usually have low current gain. On account of these factors, the design of suitable drive circuits
for BJTs is a demanding task.
Requirements of Base Drive
A good base drive circuit must satisfy the following general requirements
1. A fast rising (< 1μS) current to turn on the device fast.
2. A hard drive of adequate magnitude (IB ) to reduce turn on loss.
3. A steady base current of adequate magnitude (IB) to keep the device in saturation during the on period of the switch.
4. A fast falling (< 1μS) current of adequate magnitude (IB-) during the storage time (typically 5 to 10 S) of the turn-o period of the switch.
5. A base voltage of adequate negative magnitude (typically 5V) during the off period of the device. This base voltage that is applied during the off
period must be through a low impedance to ensure good dV/dt immunity during the off period.
6. The drive circuit must be such that the switching performance is insensitive to the operating point of the switch.
7. Electrical isolation between the control input and the switch may be desired. This will be necessary very often when the system has several
switches located at dierent electrical potentials.
8. The drive circuit must have overriding protection to switch o the device under fault.
The preferred base drive is illustrated in Fig. 3.1. The desired features of a good drive circuit are
Fast rising current for fast turn on.
Hard turn on drive to reduce turn on loss.
Adequate drive for low conduction test.
Negative base drive to reduce storage time.
Negative base bias for good dv/dt immunity. Base drive current is zero under this condition.
There are several drive circuits, which satisfy these requirements. Some of these circuits are described here.
Snubber Circuits for Power Switching Devices
Real power switching devices take a nite time to switch on or off. During the switch-on time the device voltage is dened. During switch-o the device current is dened. The second quantity during switching (device current during turn-on and device voltage during turn-o) is decided by the external circuit to the switch. In many applications, the load will be inductive (motor drives, inductive lters). The power circuit will have parasitic inductance associated with the conducting paths. Further there will be several other non-idealities of the switches present. On account of all these factors, the switching process in the device will be far from ideal. Figure 3.10 shows a simple chopper circuit consisting of all ideal components except the power switching device (in this case a BJT). The load being inductive, may be considered to be a constant current branch for the purpose of analysis. The switch voltage, current, switching energy loss and the v-i trajectory of the switch current and voltage on the vi plane in course of switching are shown in Fig. 3.11. The peak power dissipation in the device is seen to be quite large (VGIL). The switching loss will be proportional to the switching frequency and is equal to
The practical circuits will have several nonidealities as listed above. The switching loci with some of these nonidealities are shown in Fig. 3.12. The current overshoot (1) is on account of the reverse recovery current of the diode. The voltage overshoot (2) is on account of the stray inductances and capacitances in the circuit. The important point to notice is that the peak voltage and current stresses on the switching device are far more than the circuit voltage and the load current. The switching loci traverse far from the axes of the v-i plane, thus indicating large transient losses. When these loci cross the safe operating area of the v-i plane, device failure is certain. The purpose of the snubber circuits for power switching devices is to reduce the switching losses by constraining the switching trajectories to move close to the vi axes during the switching transient. From the non-ideal switching loci, it may be seen that over currents occur during turn-on and over voltage during turn-off. The snubber ensures that during turn-on rate of rise of current is limited (with a series inductor), and during turn-o the rate of rise of voltage is limited (with a shunt capacitor). The other elements in the snubber are to reduce the eects of turn-on snubber on the turn-o process and turn-off snubber on the turn-on process. The snubber circuit caters to three functions.
Turn-off aid
Turn-on aid
Over-voltage suppression
Types of Capacitors
There are several different types of capacitors employed for power electronic applications.
Coupling Capacitors
Coupling capacitors are used to transfer ac voltages between two circuits at dierent average potentials. Such capacitors are employed mostly in control circuits to couple ac signals from one circuit to another with diering dc potentials. The current carried by such a capacitor is comparatively low. The important feature of such capacitors is High insulation resistance
Power capacitors (low frequency)
These are used in PES mainly to improve power factor. They are generally used at low frequencies (predominantly 50/60 Hz). They compensate the reactive power demanded by the load so that the power handling portion of the PES are not called upon to supply the reactive power. Further, they also bypass harmonics generated in the PES. In such applications the voltage is predominantly sinusoidal; the current may be rich in harmonics. The important features of these capacitors are
Capability to handle high reactive power.
Capability to handle high harmonic current.
A typical application is shown in Fig. 2.10
Power capacitors (high frequency)
These are used for the same applications as the low frequency power capacitors but at higher frequencies (up to 20 kHz). Further they are also capable of carrying surge current resulting from switching. Such applications arise when capacitor banks are switched on and o to cater to conditions of varying load (typical in induction heating applications). The main features of these capacitors are,
Capability to handle large reactive power.
Capability to operate at higher frequency.
Capability to handle switching surge currents.
Filter capacitors
These capacitors are forward ltering capacitors to smooth out the variable source voltage applied to the load or reverse ltering capacitor to smooth out the variable load current from reaching the source. They are called upon to handle large periodic currents. The important features of these capacitors are,
High capacitors value.
High rms current rating.
These capacitors are electrolytic capacitors on account of the unipolar voltage they are subjected to. Typical applications are shown in Fig. 2.11.
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